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ISL8502
Data Sheet January 17, 2007 FN6389.0
2.5A Synchronous Buck Regulator with Integrated MOSFETs
The ISL8502 is a synchronous buck controller with internal MOSFETs packaged in a small 4mmx4mm QFN package. The ISL8502 can support a continuous load of 2.5A and has a very wide input voltage range. With the switching MOSFETs integrated into the IC, the complete regulator footprint can be very small and provide a much more efficient solution than a linear regulator. The ISL8502 is capable of stand alone operation or it can be used in a master slave combination for multiple outputs that are derived from the same input rail. Multiple slave channels (up to six) can be synchronized. This method minimizes the EMI and beat frequencies effect with multi-channel operation. The switching PWM controller drives two internal N-Channel MOSFETs in a synchronous-rectified buck converter topology. The synchronous buck converter uses voltagemode control with fast transient response. The switching regulator provides a maximum static regulation tolerance of 1% over line, load, and temperature ranges. The output is user-adjustable by means of external resistors down to 0.6V. The output is monitored for undervoltage events. The switching regulator also has overcurrent protection. Thermal shutdown is integrated. The ISL8502 features a bidirectional Enable pin that allows the part to pull the enable pin low during fault detection.
Features
* Over 2.5A Continuous Output Current * Integrated MOSFETs for Small Regulator Footprint * Adjustable Switching Frequency, 500kHz - 1.2MHz * Tight Output Voltage Regulation, 1% Over Temperature * Wide Input Voltage Range, 5V 10% or 5.5V to 14V * Wide Output Voltage Range, from 0.6V * Simple Single-Loop Voltage-Mode PWM Control Design * Input Voltage Feed-Forward for Constant Modulator Gain * Fast PWM Converter Transient Response * Lossless rDS(ON) High Side and Low Side Overcurrent Protection * Undervoltage Detection * Integrated Thermal Shutdown Protection * Power Good Indication * Adjustable Soft-Start * Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
* Point of Load Applications * Graphics Cards - GPU and Memory Supplies * ASIC Power Supplies * Embedded Processor and I/O supplies * DSP Supplies
Pinout
ISL8502 (24 LD QFN) TOP VIEW
BOOT PVCC
Ordering Information
VCC VIN VIN VIN 24 PGOOD SGND EN SYNCH M/S FS 1 2 3 4 5 6 7 COMP 8 FB 9 SS 10 PGND 11 PGND 12 PGND GND 25 23 22 21 20 19 18 VIN 17 PHASE 16 PHASE 15 PHASE 14 PHASE 13 PGND
PART NUMBER ISL8502IRZ* (Note)
PART TEMP. MARKING RANGE (C) 85 02IRZ -40 to +85
PACKAGE
PKG. DWG. #
24 Ld 4x4 QFN L24.4x4D (Pb-free)
*Add "-T" suffix for tape and reel. NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
b CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright (c) Intersil Americas Inc. 2006, 2007. All Rights Reserved All other trademarks mentioned are the property of their respective owners.
Block Diagram
PVCC
VCC
SS
PGOOD
VIN (x4)
VIN PVCC SERIES REGULATOR POR MONITOR 30A OC MONITOR
2
SGND EN SYNCH M/S FS
FN6389.0 January 17, 2007
BIAS PVCC
BOOT FAULT MONITORING
ISL8502
VOLTAGE MONITOR
GATE DRIVE AND ADAPTIVE SHOOT THRU PROTECTION
PHASE (x4)
CLOCK AND OSCILLATOR GENERATOR 0.6V REFERENCE OC MONITOR
FB
COMP
PGND (x4)
ISL8502 Typical Application Schematic
POWER GOOD ENABLE PGOOD EN SYNCH M/S VCC PVCC SS BOOT VIN VIN 5.5V to 14V
+
ISL8502
PHASE + PGND
VOUT
FS FB SGND COMP
STAND ALONE REGULATOR: VIN 5.5V TO 14V
Typical Application Schematic
VIN 4.5V TO 5.5V POWER GOOD ENABLE PGOOD EN PVCC VCC SS BOOT
VIN
+
ISL8502
PHASE +
VOUT
SYNCH M/S FS FB SGND COMP PGND
STAND ALONE REGULATOR: VIN 4.5V TO 5.5V
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FN6389.0 January 17, 2007
ISL8502 ISL8502 With Multiple Slaved Channels
VIN MASTER M/S SS PVCC VIN VOUT1 PHASE +
FS SYNCH RT EN
GND ISL8502 ENABLE M/S 5K RT FS VOUT2 SYNCH EN ISL8502 SLAVE PHASE + VIN
GND
M/S 5K RT FS
VIN VOUTN
SYNCH EN
PHASE
+
GND ISL8502 SLAVE
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FN6389.0 January 17, 2007
ISL8502
Absolute Maximum Ratings
VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +16.5V VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +6.0V Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . +22.0V Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . +6.0V All other Pins . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to VCC + 0.3V
Thermal Information
Thermal Resistance
JA (C/W) JC (C/W)
QFN Package (Notes 1, 2) . . . . . . . . . 39 2.5 Maximum Junction Temperature (Plastic Package) . . . . . . +150C Maximum Storage Temperature Range . . . . . . . . . -65C to +150C Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . +300C (SOIC - Lead Tips Only)
Recommended Operating Conditions
Supply Voltage on VIN . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5V to 14V Ambient Temperature Range . . . . . . . . . . . . . . . . . . -40C to +85C Junction Temperature Range. . . . . . . . . . . . . . . . . -40C to +125C
CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. 150C max junction temperature is intended for short periods of time to prevent shortening the lifetime. Operation close to 150C junction may trigger the shutdown of the device even before 150C, since this number is specified as typical.
NOTE: 1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with "direct attach" features. See Tech Brief TB379. 2. For JC, the "case temp" location is the center of the exposed metal pad on the package underside. 3. Not production tested. 4. Minimum VIN can operate below 5.5V as long as VCC is greater than 4.5V. 5. Maximum VIN can be higher than 14V voltage stress across the upper and lower do not exceed 15.5V in all conditions. 6. Circuit requires 100ns minimum on time to detect over current condition.
Electrical Specifications
Refer to Block and Simplified Power System Diagrams and Typical Application Schematics. Operating Conditions Unless Otherwise Noted: Vin = 12V, or VCC = 5V 10%, TA = -40C to +85C. Typical are at TA = +25C. SYMBOL TEST CONDITIONS MIN TYP MAX UNITS
PARAMETER VIN SUPPLY Input Voltage Range
VIN VIN tied to VCC
5.54 4.5
145 5.5 7 1.25 2
V V mA mA
Input Operating Supply Current Input Standby Supply Current SERIES REGULATOR VCC Voltage Maximum Output Current VCC Current Limit POWER-ON RESET Rising VCC POR Threshold Falling VCC POR Threshold ENABLE Rising Enable Threshold Voltage Falling Enable Threshold Voltage Enable Sinking Current OSCILLATOR PWM Frequency
IQ IQ_SBY
VFB = 1.0V EN tied to GND, VIN = 14V
VPVCC IPVCC
VIN > 5.6V VIN = 12V VIN = 12V, VCC shorted to PGND. Note 3
4.5 50
5.0
5.5
V mA
300
mA
4.2 3.85
4.4 4.0
4.49 4.10
V V
VEN_Rising VEN_Fall IEN
2.7 2.3 500
V V A
fOSC
RT = 96k RT = 40k FS pin tied to VCC
400 960
500 1200 800 1.0
600 1440
kHz kHz kHz V
Ramp Amplitude
VOSC
VIN = 14V
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FN6389.0 January 17, 2007
ISL8502
Electrical Specifications
Refer to Block and Simplified Power System Diagrams and Typical Application Schematics. Operating Conditions Unless Otherwise Noted: Vin = 12V, or VCC = 5V 10%, TA = -40C to +85C. Typical are at TA = +25C. (Continued) SYMBOL VOSC VVIN/VOSC DMAX DMAX VIN = 5V By Design fOSC = 500kHz fOSC = 1.2MHz 88 76 TEST CONDITIONS MIN TYP 0.470 8 MAX UNITS V % %
PARAMETER Ramp Amplitude Modulator Gain Maximum Duty Cycle Maximum Duty Cycle REFERENCE VOLTAGE Reference Voltage System Accuracy FB Pin Bias Current SOFT-START Soft-Start Current Enable Soft-Start Threshold Enable Soft-Start Threshold Hysteresis Enable Soft-Start Voltage High ERROR AMPLIFIER DC Gain Gain-Bandwidth Product Maximum Output Voltage Slew Rate INTERNAL MOSFETS Upper MOSFET rDS(ON) Lower MOSFET rDS(ON) PGOOD PGOOD Threshold
VREF -1.0
0.600 +1.0 80 200
V % nA
ISS
20 0.8
30 1.0 12
40 1.2
A V mV
2.8
3.2
3.8
V
Note 3 GBWP Note 3 3.9 SR
88 15 4.4 5
dB MHz V V/s
rDS_Upper rDS_Lower
VCC = 5V VCC = 5V
180 90
m m
VFB/VREF
Rising Edge Hysteresis 1% Falling Edge Hysteresis 1%
107 86
111 90 250
115 93
% % ms
PGOOD Rising Delay PGOOD Leakage Current PGOOD Low Voltage PGOOD Sinking Current PROTECTION Positive Current Limit Negative Current Limit Undervoltage Level Thermal Shutdown Setpoint Thermal Recovery Setpoint
tPGOOD_DELAY
fOSC = 500kHz VPGOOD = 5.5V
5 0.10 0.5
A V mA
VPGOOD IPGOOD
Note 3 Note 3
IPOC INOC VFB/VREF TSD TSR
VCC = 5V, IOC from PHASE to PGND. Notes 3, 6 VCC = 5V, IOC from VIN to PHASE. Notes 3, 6
2.8 2.2 76
3.5 3.0 80 150 130
4.0 3.5 84
A A % C C
Note 3 Note 3
6
FN6389.0 January 17, 2007
ISL8502 Typical Performance Curves
80 70 60 EFFICIENCY (%) EFFICIENCY (%) 50 40 30 20 10 0 0 1 OUTPUT LOAD (A) 2 5VIN 9VIN 14VIN
VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7H, CIN = 20F, COUT = 100F+22F, TA = +25 C, unless otherwise noted.
90 80 70 60 50 40 30 20 10 0 0 1 OUTPUT LOAD (A) 2 5VIN 9VIN
14VIN
FIGURE 1. EFFICIENCY vs LOAD (VOUT = 0.6V 500kHz)
FIGURE 2. EFFICIENCY vs LOAD (VOUT = 1.2V 500kHz)
90 80 70 EFFICIENCY (%) 14VIN EFFICIENCY (%) 60 50 40 30 20 10 0 0 1 OUTPUT LOAD (A) 2 5VIN 9VIN
90 80 70 60 50 40 30 20 10 0 0 1 OUTPUT LOAD (A) 2 5VIN 9VIN 14VIN
FIGURE 3. EFFICIENCY vs LOAD (VOUT = 1.5V 500kHz)
FIGURE 4. EFFICIENCY vs LOAD (VOUT = 1.8V 500kHz)
100 90 80 EFFICIENCY (%) EFFICIENCY (%) 70 60 50 40 30 20 10 0 0 1 OUTPUT LOAD (A) 2 5VIN 9VIN 14VIN
100 90 80 70 9VIN 60 50 40 30 20 10 0 0 1 OUTPUT LOAD (A) 2 5VIN 14VIN
FIGURE 5. EFFICIENCY vs LOAD (VOUT = 2.5V 500kHz)
FIGURE 6. EFFICIENCY vs LOAD (VOUT = 3.3V 500kHz)
7
FN6389.0 January 17, 2007
ISL8502 Typical Performance Curves
100 90 80 OUTPUT VOLTAGE (V) EFFICIENCY (%) 70 60 50 40 30 20 0.6020 10 0 0 1 OUTPUT LOAD (A) 2 0.6019 0 5VIN 1 OUTPUT LOAD (A) 2 7VIN 9VIN 14VIN 0.6024 0.6023 14VIN 0.6022 0.6021
VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7H, CIN = 20F, COUT = 100F+22F, TA = +25 C, unless otherwise noted. (Continued)
0.6026 0.6025
9VIN
FIGURE 7. EFFICIENCY vs LOAD (5V 500kHz)
FIGURE 8. VOUT REGULATION vs LOAD (VOUT = 0.6V 500kHz)
1.206 14VIN 1.205 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V)
1.520
1.518 5VIN 1.516
1.204 9VIN
1.203
1.514 9VIN 1.512 14VIN
1.202 5VIN 1.201
1.200
0
1 OUTPUT LOAD (A)
2
1.510
0
1 OUTPUT LOAD (A)
2
FIGURE 9. VOUT REGULATION vs LOAD (VOUT = 1.2V 500kHz)
FIGURE 10. VOUT REGULATION vs LOAD (VOUT = 1.5V 500kHz)
1.815 1.815 1.814 OUTPUT VOLTAGE (V) 1.814 1.813 1.813 1.812 1.812 1.811 1.811 1.810 0 1 OUTPUT LOAD (A) 2 14VIN 9VIN 5VIN
2.515
2.513 OUTPUT VOLTAGE (V) 14VIN 2.511
2.509 5VIN 2.507 9VIN
2.505
0
1 OUTPUT LOAD (A)
2
FIGURE 11. VOUT REGULATION vs LOAD (VOUT = 1.8V 500kHz)
FIGURE 12. VOUT REGULATION vs LOAD (VOUT = 2.5V 500kHz)
8
FN6389.0 January 17, 2007
ISL8502 Typical Performance Curves
3.355 3.354 3.353 OUTPUT VOLTAGE (V) 3.352 3.351 3.350 3.349 3.348 3.347 3.346 3.345 0 1 OUTPUT LOAD (A) 2 9VIN 9VIN 5.020 0 1 OUTPUT LOAD (A) 2 14VIN OUTPUT VOLTAGE (V) 5VIN 5.028 7VIN 5.026
VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7H, CIN = 20F, COUT = 100F+22F, TA = +25 C, unless otherwise noted. (Continued)
5.030
5.024
5.022
14VIN
FIGURE 13. VOUT REGULATION vs LOAD (VOUT = 3.3V 500kHz)
2.0 1.8 POWER DISSIPATION (W)
FIGURE 14. VOUT REGULATION vs LOAD (VOUT = 5V 500kHz)
2.0 1.8 1.6 POWER DISSIPATION (W) 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 9VIN 0 1 OUTPUT LOAD (A) 2 14VIN 5VIN
1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 0 1 OUTPUT LOAD (A) 2 9VIN 5VIN 14VIN
FIGURE 15. POWER DISSIPATION vs LOAD (VOUT = 0.6V 500kHz)
FIGURE 16. POWER DISSIPATION vs LOAD (VOUT = 1.2V 500kHz)
2.5
2.5
2.0 POWER DISSIPATION (W) POWER DISSIPATION (W)
2.0
1.5
1.5 14VIN 1.0
1.0 14VIN 0.5 9VIN 0.0
5VIN
0.5 9VIN 0.0
5VIN
0
1 OUTPUT LOAD (A)
2
0
1 OUTPUT LOAD (A)
2
FIGURE 17. POWER DISSIPATION vs LOAD (VOUT = 1.5V 500kHz)
FIGURE 18. POWER DISSIPATION vs LOAD (VOUT = 1.8V 500kHz)
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FN6389.0 January 17, 2007
ISL8502 Typical Performance Curves
2.5
VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7H, CIN = 20F, COUT = 100F+22F, TA = +25 C, unless otherwise noted. (Continued)
2.5
POWER DISSIPATION (W)
POWER DISSIPATION (W)
2.0
2.0 14VIN 1.5
1.5 14VIN 1.0
1.0
0.5 9VIN 0 1 OUTPUT LOAD (A)
5VIN
0.5 5VIN
0.0
0.0 2 0
9VIN 1 OUTPUT LOAD (A) 2
FIGURE 19. POWER DISSIPATION vs LOAD (VOUT = 2.5V 500kHz)
FIGURE 20. POWER DISSIPATION vs LOAD (VOUT = 3.3V 500kHz)
2.5
5.2 5.1 5.0
POWER DISSIPATION (W)
2.0
1.5
VCC (V) 7VIN 9VIN
14VIN
4.9 4.8 4.7 4.6 4.5
1.0
0.5
0.0 0 1 OUTPUT LOAD (A) 2
0
50
100
150 I VCC (mA)
200
250
300
FIGURE 21. POWER DISSIPATION vs LOAD (VOUT = 5V 500kHz)
5.5 5.4 5.3 5.2 VCC (V) 5.1 5.0 4.9 4.8 4.7 4.6 4.5 3 4 5 6 7 8 9 10 VIN (V) 11 12 13 14 15 100mA LOAD NO LOAD
FIGURE 22. VCC LOAD REGULATION
PHASE1 0.5s 5V/DIV 5V PHASE2 5V/DIV
VOUT1 RIPPLE 20mV/DIV
VOUT2 RIPPLE 20mV/DIV
FIGURE 23. VCC REGULATION vs VIN
FIGURE 24. MASTER TO SLAVE OPERATION
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FN6389.0 January 17, 2007
ISL8502 Typical Performance Curves
VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7H, CIN = 20F, COUT = 100F+22F, TA = +25 C, unless otherwise noted. (Continued)
PHASE1 5V/DIV
PHASE1 5V/DIV
VOUT1 RIPPLE 20mV/DIV
VOUT1 RIPPLE 20mV/DIV IL1 1A/DIV
IL1 0.5A/DIV SYNCH1 5V/DIV SYNCH1 2V/DIV
FIGURE 25. MASTER OPERATION AT NO LOAD
FIGURE 26. MASTER OPERATION WITH FULL LOAD
PHASE1 10V/DIV
EN1 5V/DIV VOUT1 1V/DIV
VOUT1 RIPPLE 20mV/DIV
IL1 1A/DIV SYNCH1 5V/DIV
IL1 2A/DIV
SS1 2V/DIV
FIGURE 27. MASTER OPERATION WITH NEGATIVE LOAD
FIGURE 28. SOFT-START AT NO LOAD
EN1 5V/DIV
PHASE1 10V/DIV
VOUT1 1V/DIV IL1 1A/DIV IL1 1A/DIV VOUT1 1V/DIV
SS1 2V/DIV
PGOOD1 5V/DIV
FIGURE 29. SOFT-START AT FULL LOAD
FIGURE 30. POSITIVE OUTPUT SHORT CIRCUIT
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FN6389.0 January 17, 2007
ISL8502 Typical Performance Curves
VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7H, CIN = 20F, COUT = 100F+22F, TA = +25 C, unless otherwise noted. (Continued)
PHASE1 10V/DIV PHASE1 10V/DIV
VOUT1 2V/DIV
VOUT1 2V/DIV IL1 2A/DIV
IL1 2A/DIV
SS1 2V/DIV
PGOOD1 5V/DIV
FIGURE 31. POSITIVE OUTPUT SHORT CIRCUIT (HICCUP MODE)
FIGURE 32. NEGATIVE OUTPUT SHORT CIRCUIT
PHASE1 10V/DIV
VOUT1 1V/DIV
PHASE1 5V/DIV VOUT1 RIPPLE 50mV/DIV
IL1 1A/DIV
IL1 2A/DIV PGOOD1 5V/DIV IOUT1 2A/DIV
FIGURE 33. RECOVER FROM POSITIVE SHORT CIRCUIT
FIGURE 34. LOAD TRANSIENT
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com 12
FN6389.0 January 17, 2007
ISL8502 Functional Pin Description
PGOOD (Pin 1)
PGOOD is an open drain output that will pull to low if the output goes out of regulation or a fault is detected. PGOOD is equipped with a fixed delay upon output power-up. The delay is approximately 250ms at switching frequency 500kHz and 108ms at 1.2MHz.
COMP (Pin 7) and FB (Pin 8)
The switching regulator employs a single voltage control loop. FB is the negative input to the voltage loop error amplifier. The output voltage is set by an external resistor divider connected to FB. With a properly selected divider, the output voltage can be set to any voltage between the power rail (reduced by converter losses) and the 0.6V reference. Loop compensation is achieved by connecting an AC network across COMP and FB. The FB pin is also monitored for undervoltage events.
SGND (Pin 2)
The SGND terminal of the ISL8502 provides the return path for the control and monitor portions of the IC.
EN (Pin 3)
The Enable pin is a bidirectional pin. If the voltage on this pin exceeds the enable threshold voltage, the part is enabled. If a fault is detected, the EN pin will be pulled low via internal circuitry for a duration of 4 soft-start periods. For automatic start-up, use 10k to 100k pull up resistor connecting to VCC.
SS (Pin 9)
Connect a capacitor from this pin to ground. This capacitor, along with an internal 30A current source, sets the soft-start interval of the converter, TSS.
C SS [ F ] = 50 T SS [ S ] (EQ. 2)
SYNCH (Pin 4)
This is a bidirectional pin that is used to synchronize slave devices to the Master device. As a Master device, this pin outputs the clock signal that the slave devices use to synchronize to. As a slave device, this pin is an input to receive the clock signal from the master device. If configured as a slave device, the ISL8502 will be disabled if there is no clock signal from the master device on the SYNCH pin. Leave this pin unconnected if the IC is used in stand alone operation.
PGND (Pins 10-13)
These pins are used as the ground connection of the power train.
PHASE (Pins 14-17)
These pins are the PHASE node connections to the inductor. These pins are connected the source of the control MOSFET and the drain of the synchronous MOSFET.
VIN (Pins 18-21)
Connect the input rail to these pins. These pins are the input to the regulator as well as the source for the internal linear regulator that supplies the bias for the IC. It is recommended that the DC voltage applied to the VIN pins does not exceed 14V. This recommendation allows for transient spikes and voltage ringing to occur while not exceeding Absolute Maximum Ratings.
M/S (Pin 5)
As a slave device, tie a 5k resistor between this pin and ground. As a master or a stand alone device, tie this pin directly to the VCC pin. Do not short M/S pin to GND.
BOOT (Pin 22)
This pin provides ground referenced bias voltage to the upper MOSFET driver. A bootstrap circuit is used to create a voltage suitable to drive the internal N-channel MOSFET. The boot diode is included within the ISL8502.
FS (Pin 6)
This pin provides oscillator switching frequency adjustment. By placing a resistor (RT) from this pin to GND, the switching frequency can be programmed as desired between 500kHz and 1.2MHz.
48000 R T [ k ] = ----------------------------f OSC [ kHz ]
PVCC (Pin 23)
This pin is the output of the internal linear regulator that supplies the bias and gate voltage for the IC. A minimum 4.7F decoupling capacitor is recommended.
(EQ. 1)
Tying the FS pin to the VCC pin will force the switching frequency to 800kHz. Using resistors with values below 40k (1.2MHz) or with values higher than 97k (500kHz) may damage the ISL8502.
VCC (Pin 24)
This pin supplies the bias voltage for the IC. This pin should be tied to the PVCC pin through an RC low pass filter. A 10 resistor and 0.1F capacitor is recommended.
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FN6389.0 January 17, 2007
ISL8502 Functional Description
Initialization
The ISL8502 automatically initializes upon receipt of input power. The Power-On Reset (POR) function continually monitors the voltage on the VCC pin. If the voltage on the EN pin exceeds its rising threshold, then the POR function initiates soft-start operation after the bias voltage has exceeded the POR threshold. If the Master device is disabled via the EN pin, it will continue to send the clock signal from the SYNCH pin. This allows slave devices to continue operation.
Fault Protection
The ISL8502 monitors the output of the regulator for overcurrent and undervoltage events. The ISL8502 also provides protection from excessive junction temperatures. Overcurrent Protection The overcurrent function protects the switching converter from a shorted output by monitoring the current flowing through both the upper and lower MOSFETs. Upon detection of any overcurrent condition, the upper MOSFET will be immediately turned off and will not be turned on again until the next switching cycle. Upon detection of the initial overcurrent condition, the Overcurrent Fault Counter is set to 1 and the Overcurrent Condition Flag is set from LOW to HIGH. If, on the subsequent cycle, another overcurrent condition is detected, the OC Fault Counter will be incremented. If there are eight sequential OC fault detections, the regulator will be shut down under an Overcurrent Fault Condition and the EN pin will be pulled LOW. An Overcurrent Fault Condition will result with the regulator attempting to restart in a hiccup mode with the delay between restarts being 4 soft-start periods. At the end of the fourth soft-start wait period, the fault counters are reset, the EN pin is released, and soft-start is attempted again. If the overcurrent condition goes away prior to the OC Fault Counter reaching a count of four, the Overcurrent Condition Flag will set back to LOW. If the Overcurrent Condition Flag is HIGH and the Overcurrent Fault Counter is less than four and an undervoltage event is detected, the regulator will be shut down immediately. Undervoltage Protection If the voltage detected on the FB pin falls 18% below the internal reference voltage and the overcurrent condition flag is LOW, then the regulator will be shutdown immediately under an Undervoltage Fault Condition and the EN pin will be pulled LOW. An Undervoltage Fault Condition will result with the regulator attempting to restart in a hiccup mode with the delay between restarts being 4 soft-start periods. At the end of the fourth soft-start wait period, the fault counters are reset, the EN pin is released, and soft-start is attempted again. Thermal Protection If the ISL8502 IC junction temperature reaches a nominal temperature of +140C, the regulator will be disabled. The ISL8502 will not re-enable the regulator until the junction temperature drops below +110C.
Stand Alone Operation
The ISL8502 can be configured to function as a stand alone single channel voltage mode synchronous buck PWM voltage regulator. The Typical Power Diagrams on Page 3 show the two configurations for stand alone operation. The internal series linear regulator requires at least 5.5V to create the proper bias for the IC. If the input voltage is between 5.5V and 15V, simply connect the VIN pins to the input rail and the series linear regulator will create the bias for the IC. The VCC pin should be tied to a capacitor for decoupling. If the input voltage is 5V 10%, then tie the VIN pins and the VCC pin to the input rail. The ISL8502 will use the 5V rail as the bias. A decoupling capacitor should be placed as close as possible to the VCC pin.
Multi-Channel (Master/Slave) Operation
The ISL8502 can be configured to function in a multichannel system. The Typical Power Diagram on Page 4 shows a typical configuration for the multi-channel system. In the multi-channel system, each ISL8502 IC regulates a separate rail while sharing the same input rail. By configuring the devices in a master/slave configuration, the clocks of each IC can be synchronized. There can only be one master IC in a multi-channel system. To configure an IC as the master, the M/S pin must be shorted to the VCC pin. The SYNCH pins of all the ISL8502 controller ICs in the multi-channel system must be tied together. The frequency set resistor value (RT) used on the master device must be used on every slave device. Each slave device must have a 5k resistor connecting it from M/S pin to ground. The master device and all the slave devices can have their EN pins tied to an enable `bus'. Since the EN pin is bidirectional, this allows for options on how each IC is tied to the enable `bus'. If the EN pin of any ISL8502 is tied directly to the enable bus, then that device will be capable of disabling all the other devices that have their EN pins tied directly to the enable bus. If the EN pin of an ISL8502 is tied to the enable bus through a diode (anode tied to ISL8502 EN pin, cathode tied to enable bus) then this part will not disable other devices on the enable bus if it disables itself for any reason.
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Shoot-Through Protection A shoot-through condition occurs when both the upper and lower MOSFETs are turned on simultaneously, effectively shorting the input voltage to ground. To protect from a shootthrough condition, the ISL8502 incorporates specialized circuitry which insures that the complementary MOSFETs are not ON simultaneously.
VOUT
VHUMP
VESR VSAG VESL
Application Guidelines
Operating Frequency
The ISL8502 can operate at switching frequencies from 500kHz to 1.2MHz. A resistor tied from the FS pin to ground is used to program the switching frequency through the following Equation 3.
48000 R T [ k ] = ----------------------------f OSC [ kHz ]
IOUT
Itran
(EQ. 3)
If the FS pin is left unconnected, the ISL8502 will default to a 500kHz switching frequency.
FIGURE 35. TYPICAL TRANSIENT RESPONSE
Output Voltage Selection
The output voltage of the regulator can be programmed via an external resistor divider that is used to scale the output voltage relative to the internal reference voltage and feed it back to the inverting input of the error amplifier. Refer to Figure 36. The output voltage programming resistor, R4, will depend on the value chosen for the feedback resistor and the desired output voltage of the regulator. The value for the feedback resistor is typically between 1k and 10k.
R 1 x 0.6V R 4 = ---------------------------------V OUT - 0.6V (EQ. 4)
High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. The shape of the output voltage waveform during a load transient that represents the worst case loading conditions will ultimately determine the number of output capacitors and their type. When this load transient is applied to the converter, most of the energy required by the load is initially delivered from the output capacitors. This is due to the finite amount of time required for the inductor current to slew up to the level of the output current required by the load. This phenomenon results in a temporary dip in the output voltage. At the very edge of the transient, the Equivalent Series Inductance (ESL) of each capacitor induces a spike that adds on top of the existing voltage drop due to the Equivalent Series Resistance (ESR). After the initial spike, attributable to the ESR and ESL of the capacitors, the output voltage experiences sag. This sag is a direct consequence of the amount of capacitance on the output. During the removal of the same output load, the energy stored in the inductor is dumped into the output capacitors. This energy dumping creates a temporary hump in the output voltage. This hump, as with the sag, can be attributed to the total amount of capacitance on the output. Figure 35 shows a typical response to a load transient.
If the output voltage desired is 0.6V, then R4 is left unpopulated.
Output Capacitor Selection
An output capacitor is required to filter the inductor current and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (Effective Series Resistance) and voltage rating requirements rather than actual capacitance requirements.
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The amplitudes of the different types of voltage excursions can be approximated using Equation 5.
V ESR = ESR * I tran
2
dI tran V ESL = ESL * --------------dt
Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter's response time to a load transient. One of the parameters limiting the converter's response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL8502 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. Equation 9 gives the approximate response time interval for application and removal of a transient load:
tRISE = L x ITRAN VIN - VOUT tFALL = L x ITRAN VOUT (EQ. 9)
L out * I tran V SAG = ------------------------------------------------C out * ( V in - V out ) L out * I tran V HUMP = ------------------------------C out * V out
2
(EQ. 5)
where: Itran = Output Load Current Transient and Cout = Total Output Capacitance In a typical converter design, the ESR of the output capacitor bank dominates the transient response. The ESR and the ESL are typically the major contributing factors in determining the output capacitance. The number of output capacitors can be determined by using Equation 6, which relates the ESR and ESL of the capacitors to the transient load step and the voltage limit (DVo):
ESL * dI tran --------------------------------- + ESR * I tran dt Number of Caps = ----------------------------------------------------------------------V o
(EQ. 6)
If DVSAG and/or DVHUMP are found to be too large for the output voltage limits, then the amount of capacitance may need to be increased. In this situation, a trade off between output inductance and output capacitance may be necessary. The ESL of the capacitors, which is an important parameter in the above equations, is not usually listed in databooks. Practically, it can be approximated using Equation 7 if an Impedance Vs. Frequency curve is given for a specific capacitor:
1 ESL = ---------------------------------------2 C ( 2 * * f res )
where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. The worst case response time can be either at the application or removal of load. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time the upper MOSFET turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of the upper MOSFET and the source of the lower MOSFET. The important parameters for the bulk input capacitance are the voltage rating and the RMS current rating. For reliable operation, select bulk capacitors with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. Their voltage rating should be at least 1.25 times greater than the maximum input voltage, while a voltage rating of 1.5 times is a conservative guideline. For most cases, the RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. The maximum RMS current through the input capacitors may be closely approximated using Equation 10:
2 V OUT V OUT 2 1 V IN - V OUT V OUT ------------- x I OUT x 1 - ------------- + ----- x ---------------------------- x ------------- 12 L x f V IN V IN V IN MAX OSC
(EQ. 7)
where: fres is the frequency where the lowest impedance is achieved (resonant frequency). The ESL of the capacitors becomes a concern when designing circuits that supply power to loads with high rates of change in the current.
Output Inductor Selection
The output inductor is selected to meet the output voltage ripple requirements and minimize the converter's response time to the load transient. The inductor value determines the converter's ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by using Equation 8:
I = VIN - VOUT Fs x L x VOUT VIN VOUT = I x ESR (EQ. 8)
(EQ. 10)
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For a through hole design, several electrolytic capacitors may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. Some capacitor series available from reputable manufacturers are surge current tested.
Modulator Break Frequency Equations
1 F LC = -----------------------------------------2 x L O x C O 1 F ESR = ------------------------------------------2 x ESR x C O (EQ. 11)
Feedback Compensation
Figure 36 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). The modulator transfer function is the small-signal transfer function of VOUT/VE/A . This function is dominated by a DC Gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR . The DC Gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage DVOSC . The ISL8502 incorporates a feed forward loop that accounts for changes in the input voltage. This maintains a constant modulator gain.
OSC PWM COMPARATOR VOSC DRIVER VIN LO DRIVER PHASE CO VOUT
The compensation network consists of the error amplifier (internal to the ISL8502) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. Equation 12 below relates the compensation network's poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figure 38. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth. 2. Place 1ST Zero Below Filter's Double Pole (~75% FLC). 3. Place 2ND Zero at Filter's Double Pole. 4. Place 1ST Pole at the ESR Zero. 5. Place 2ND Pole at Half the Switching Frequency. 6. Check Gain against Error Amplifier's Open-Loop Gain. 7. Estimate Phase Margin - Repeat if Necessary.
Compensation Break Frequency Equations
1 F Z1 = ----------------------------------2 x R 2 x C 1 1 F Z2 = -----------------------------------------------------2 x ( R 1 + R 3 ) x C 3 1 F P1 = ------------------------------------------------------- C 1 x C 2 2 x R 2 x --------------------- C1 + C2 1 F P2 = ----------------------------------2 x R 3 x C 3 (EQ. 12)
-
+
ZFB VE/A +
ESR (PARASITIC)
-
ZIN REFERENCE
ERROR AMP
DETAILED COMPENSATION COMPONENTS C1 C2 R2 ZFB ZIN C3 R1 FB R4 R3 VOUT
COMP
Figure 37 shows an asymptotic plot of the DC/DC converter's gain vs. frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 37. Using the above guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the graph of Figure 37 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than +45. Include worst case component variations when determining phase margin. A more detailed explanation of voltage mode control of a buck regulator can be found in Tech Brief TB417, titled "Designing Stable Compensation Networks for Single Phase Voltage Mode Buck Regulators."
+
ISL8502
REFERENCE
R 1 V OUT = 0.6 x 1 + ------ R 4 FIGURE 36. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN AND OUTPUT VOLTAGE SELECTION
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In order to dissipate heat generated by the internal VTT LDO, the ground pad, pin 29, should be connected to the internal ground plane through at least five vias. This allows the heat to move away from the IC and also ties the pad to the ground plane through a low impedance path. The switching components should be placed close to the ISL8502 first. Minimize the length of the connections between the input capacitors, CIN, and the power switches by placing them nearby. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible. Position the output inductor and output capacitors between the upper and lower MOSFETs and the load. Make the PGND and the output capacitors as short as possible. The critical small signal components include any bypass capacitors, feedback components, and compensation components. Place the PWM converter compensation components close to the FB and COMP pins. The feedback resistors should be located as close as possible to the FB pin with vias tied straight to the ground plane as required.
VIN 5V CBP1 RBP VCC CBP2 PHASE PGND COMP C2 R2 FB R4 GND PAD C3 R3 C1 R1 COUT1 LOAD
FN6389.0 January 17, 2007
100 80 60 GAIN (dB) 40 20 0 -20 -40 MODULATOR GAIN 20LOG (R2/R1)
FZ1 FZ2
FP1
FP2 OPEN LOOP ERROR AMP GAIN
20LOG (VIN/VOSC) COMPENSATION GAIN CLOSED LOOP GAIN FESR 10K 100K 1M 10M
FLC -60 10 100 1K
FREQUENCY (Hz)
FIGURE 37. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Layout Considerations
Layout is very important in high frequency switching converter design. With power devices switching efficiently between 500kHz and 1.2MHz, the resulting current transitions from one device to another cause voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device overvoltage stress. Careful component layout and printed circuit board design minimizes these voltage spikes. As an example, consider the turn-off transition of the control MOSFET. Prior to turn-off, the MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is picked up by the lower MOSFET. Any parasitic inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide traces minimizes the magnitude of voltage spikes. There are two sets of critical components in the ISL8502 switching converter. The switching components are the most critical because they switch large amounts of energy, and therefore tend to generate large amounts of noise. Next are the small signal components which connect to sensitive nodes or supply critical bypass current and signal coupling. A multi-layer printed circuit board is recommended. Figure 38 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually a middle layer of the PC board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminals to the output inductor short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the GATE pins to the MOSFET gates should be kept short and wide enough to easily handle the 1A of drive current. 18
PVCC
VIN CIN ISL8502 L1 VOUT1
KEY ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT AND/OR POWER PLANE LAYER VIA CONNECTION TO GROUND PLANE
FIGURE 38. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
ISL8502
Package Outline Drawing
L24.4x4D
24 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 2, 10/06
4X 2.5 4.00 A B 19 20X 0.50 24 PIN #1 CORNER (C 0 . 25)
PIN 1 INDEX AREA
18
1
4.00
2 . 50 0 . 15
13
(4X)
0.15 12 7 0.10 M C A B 0 . 07 24X 0 . 23 + 0 . 05 4 24X 0 . 4 0 . 1
TOP VIEW
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 90 0 . 1
( 3 . 8 TYP )
C BASE PLANE
SIDE VIEW
SEATING PLANE 0.08 C
(
2 . 50 ) ( 20X 0 . 5 )
C ( 24X 0 . 25 ) ( 24X 0 . 6 )
0 . 2 REF
5
0 . 00 MIN. 0 . 05 MAX.
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 indentifier may be either a mold or mark feature.
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